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How to suppress chaotic operation in a DCM flyback at low load

I would like to share these tips with everybody.
A current mode controlled flyback converter always becomes unstable at low load due to the unavoidable leading edge current spike. It is not normally dangerous but as a design engineer I don’t like to look at it and listen to it.

Here are three useful and not patented tips.

First tip:
• Insert a low pass filter, say 1kohm + 100pF between current sense resistor and CS input in your control IC.
• Split the 1kohm in two resistors R1 to the fet and R2 to the control IC. R1 << R2.
• Insert 0,5 – 1pF between drain and the junction R1/R2. This can be made as a layer-to-layer capacitor in the PCB. It does not have to be a specific value.
• Adjust R1 until the spike in the junction in R1/R2 is cancelled.
You will see that the current spike is always proportional to the negative drain voltage step at turn-on. Once adjusted, the cancellation always follows the voltage step, and you some times achieve miracles with it. Cost = one resistor.

Second tip:
Having the low pass filter from first tip, add a small fraction of the gate driver output voltage to the current sense input, say 0,1V by inserting a large resistor from ‘Drive Out’ to ‘CS input’. The added low pass filtered step voltage will more or less conceal the current spike. You should reduce your current sense resistor accordingly. Cost = one resistor.

Third tip:
In a low power flyback, you some times just need an RC network or just an extra capacitor from drain to a DC point, either to reduce overshoot or to reduce noise. Connect the RC network or the capacitor to source, not to ground or Vcc. If you connect it to ground or Vcc, you will measure the added discharge current peak in the current sense resistor. Cost = nothing – just knowledge.

All tips can be used individually or combined => Less need for pre-load resistors on your output.

Right Half Plane Pole

Very few know about the Right Half Plane Pole (not a RHP-Zero) at high duty cycle in a DCM buck with current mode control. Maybe because it is not really a problem.
It is said that this instability starts above 2/3 duty cycle – I think that must be with a resistive load. If loaded with a pure current source, it starts above 50% duty cycle.

Here is a little down-to-earth explanation:
If you run a buck converter at high duty cycle but DCM, it probably works fine and is completely stable. Then imagine you suddenly open the feedback loop, leaving the peak current constant and unchanged. The duty cycle will then rush either back to 50% or to 100% if possible. You now have a system with a negative output resistance – if Voltage goes up, the output current will increase.

You can see it by drawing some triangles on a piece of paper: A steady state DCM current triangle with an up-slope longer than the down-slope and a fixed peak value. Now, if you imagine that the output voltage rises, you can draw a new triangle with the same peak current. The up-slope will be longer, the down-slope will be shorter but the sum of times will be longer than in the steady state case. The new triangle therefore has a larger area than the steady state triangle, which means a higher average output current. So higher output voltage generates higher output current if peak current is constant. Loaded with a current source, it is clear that this is an unstable system, like a flipflop, and it starts becoming unstable above 50% duty cycle.

However, when you close the feedback loop, the system is (conditionally) stable and the loop gain is normally so high at the RHP Pole frequency that it requires a huge gain reduction to make it unstable.

It’s like when you drive on your bike. A bike has two wheels and therefore can tilt to either side – it is a system with a low frequency RHPP like a flipflop. If you stand still, it will certainly tilt to the left or to the right because you have no way to adjust your balance back. But if you drive, you have a system with feedback where you can immediately correct imbalance by turning the handlebars. As we know, this system is stable unless you have drunk a lot of beers.

Experience: Flyback

My first SMPS design was a multiple output flyback. This was in 1976, when there were no PWM controllers. So I used a 556 (1/2 osc -30 kHz, and 1/2 PWM generator) plus used a 3904 NPN where the VBE was the reference and also provided gain for the error amp function. I hap-hazardly wound the windings on a 25 mm torroid. It ranglike a tank circuit. I quickly abandoned the transformer and after a year, and many hours on the bench, I had a production-grad SMPS.
Since it went into a private aircraft weather reader system, I needed an exterier SMPS which was a buck converter. I used an LM105 linear regulator with positive feedback to make it oscillate (one of nationals ap notes). It worked, but I soon learned that the electrolytic capacitors lost all of their capacitance at -25 deg C. It later worked with military-grade capacitors.

I had small hills of dead MOSFETs and the directly attached controllers. When the first power MOSFETs emerged in 1979, I blew-up so many that I almost wrote them off. They had some real issues with D-S voltage overstress. They have come a long way since.

As far as very wide range flyback converter, please dig-up AN1327 on the ONSEMI website. This describes a control strategy (fixed off-time, variable on-time) and the transformer design.
The processor to that was a 3W flyback that drove 3 floating gate drive circuits and had an input range of 85 VAC to 576 VAC. It was for a 3 phase industrial motor drive. The toughest area was the transformer. To meet the isolation requirements of the UL, and IEC, it would have required a very large core, and bobbin plus a lot of tape. The PCB had the dimensions of 50 mm x 50 mm and 9 mm thick A magnetics designer named Jeff Brown from Cramerco.com is now my magnetics God. He designed me a custom core and bobbin that was 10 mm high on basically an EF15 sized core. The 3 piece bobbin met all of the spacing requirements without tape. The customer was expecting a 2 – 3 tier product offering for the different voltage ranges, but instead could offer only one. They were thrilled.

Can be done, watch your breakdown voltages, spacings and RMS currents. I found that around 17 -20 watts is about the practical limit for an EF40 core before the transformer RMS currents get too high.

Experience: Design

I tell customers that at least 50% of the design effort is the layout and routing by someone who knows what they are doing. Layer stackup is very critical for multiple layer designs. Yes, a solid design is required. But the perfect design goes down in flames with a bad layout. Rudy Severns said it best in one of his early books that you have to “think RF” when doing a layout. I have followed this philosophy for years with great success. Problems with a layout person who wants to run the auto route or doesn’t understand analog layout? No problem, you, as the design engineer, do not have to release the design until it is to your satisfaction.

I have had Schottky diodes fail because the PIV was exceeded due to circuit inductance causing just enough of a very high frequency ring (very hard to see on a scope) to exceed the PIV. Know your circuit Z’s, keep your traces short and fat.

Fixed a number of problems associated with capacitor RMS ratings on AC to DC front ends. Along with this is the peak inrush current for a bridge rectifier at turn on and, in some cases, during steady state. Unit can be turned on at the 90 deg phase angle into a capacitive load. This must be analyzed with assumptions for input resistance and/or a current inrush circuit must be added.

A satellite power supply had 70 deg phase margin on the bench, resistive load, but oscillated on the real load. Measured the loop using the AP200 on the load and the phase margin was zero. Test the power supply on the real load before going to production and then a random sampling during the life of the product.

I used MathCAD for designs until the average models came out for SMPS. Yes, the equations are nice to see and work with but they are just models none the less. I would rather have PsPice to the math while I pay attention to the models used and the overall design effort. Creating large closed form equations is wrought with pitfalls, trapdoors, and landmines. Plus, hundreds of pages of MathCAD, which I have done, is hard to sell to the customer during a design review (most attendees drift off after page 1). The PsPice schematics are more easily sold and then modified as needed with better understanding all around.

Power supply prototypes is the best way to learn it

I have been designing power supplies for over 15 years now. We do mostly off line custom designs ranging from 50 to 500W. Often used in demanding environments such as offshore and shipping.
I think we are the lucky ones who got the chance to learn designing power supplies using the simple topologies like a flyback or a forward converter. If we wanted to make something fancy we used a push-pull or a half bridge.

Nowadays, straight out of school you get to work on a resonant converter, working with variable frequency control. Frequencies are driven up above 250kHz to make it fit in a matchbox, still delivering 100W or more. PCB layouts get almost impossible to make if you also have to think about costs and manufacturability.
Now the digital controllers are coming into fashion. These software designers know very little about power electronics and think they can solve every problem with a few lines of code.

But I still think the best way to learn is to start at the basics and do some through testing on the prototypes you make. In my department we have a standard test program to check if the prototype functions according to the specifications (Design Verification Tests), but also if all parts are used within their specifications (Engineering Verification Tests). These tests are done at the limits of input voltage range and output power. And be aware that the limit of the output power is not just maximum load, but also overload, short circuit and zero load! Start-up and stability are tested at low temperature and high temperature.

With today’s controllers the datasheets seem to get ever more limited in information, and the support you get from the FAE’s is often very disappointing. Sometime ago I even had one in the lab who sat next to me for half a day to solve a mysterious blow up of a high side driver. At the end of the day he thanked me, saying he had learned a lot!
Not the result I was hoping for.

OPC drivers advantage

A few years back, I had a devil of time getting some OPC Modbus TCP drivers to work with Modbus RTU to TCP converts. The OPC drivers could not handle the 5 digit RTU addressing. You need to make sure your OPC driver that you try actually works with your equipment. Try before you buy is definite here. Along with some of the complications, like dropping connections due minor network cliches, a real headache and worth a topic all its own, is the ability us tag pickers and the like. The best thing to happen to I/O addressing is the use of Data Objects in the PLC and HMI/SCADA. The other advantage OPC can give you the ability to get more Quality Information on your I/O. Again, check before you buy. In my experience, the only protocol worse than Modbus in the Quality Info department is DDE and that pretty well gone. This still does not help when the Modbus slave still reports stale data like its fresh. No I/O driver can sort that out, you need a heartbeat.

A shout out to all you Equipment manufactures that putting Modbus RTU into equipment because its easy, PLEASE BUILD IN A HEATBEAT us integrators can monitor so we can be sure the data is alive and well.

Also, while you try before you buy, you want your HMI/SCADA to be able to tell the difference between, Good Read, No Read and Bad Read, particularly with a RTU network.

High voltage power delivery

You already know from your engineering that higher voltages results to less operational losses for the same amount of power delivered. The bulk capacity of 3000MW has a great influence on the investment costs obviously, that determines the voltage level and the required number of parallel circuit. The need for higher voltage DC levels has become more feasible for bulk power projects (such as this one) especially when the transmission line is more than 1000 km long. So on the economics, investment for 800kV DC systems have been much lower since the 90’s. Aside from reduction of overall project costs, HVDC transmission lines at higher voltage levels require lesser right-of-way. Since you will be also requiring less towers as will see below, then you will also reduce the duration of the project (at least on the line).

Why DC not AC? From a technical point of view, there are no special obstacles against higher DC voltages. Maintaining stable transmission could be difficult over long AC transmission lines. The thermal loading capability is usually not decisive for long AC transmission lines due to limitations in the reactive power consumption. The power transmission capacity of HVDC lines is mainly limited by the maximum allowable conductor temperature in normal operation. However, the converter station cost is expensive and will offset the gain in reduced cost of the transmission line. Thus a short line is cheaper with ac transmission, while a longer line is cheaper with dc.
One criterion to be considered is the insulation performance which is determined by the overvoltage levels, the air clearances, the environmental conditions and the selection of insulators. The requirements on the insulation performance affect mainly the investment costs for the towers.

For the line insulation, air clearance requirements are more critical with EHVAC due to the nonlinear behavior of the switching overvoltage withstand. The air clearance requirement is a very important factor for the mechanical design of the tower. The mechanical load on the tower is considerably lower with HVDC due to less number of sub-conductors required to fulfill the corona noise limits. Corona rings will be always significantly smaller for DC than for AC due to the lack of capacitive voltage grading of DC insulators.

With EHVAC, the switching overvoltage level is the decisive parameter. Typical required air clearances at different system voltages for a range of switching overvoltage levels between 1.8 and 2.6 p.u. of the phase-to-ground peak voltage. With HVDC, the switching overvoltages are lower, in the range 1.6 to 1.8 p.u., and the air clearance is often determined by the required lightning performance of the line.

How generator designers determine the power factor?

The generator designers will have to determine the winding cross section area and specific current/mm² to satisfy the required current, and they will have to determine the required total flux and flux variation per unit of time per winding to satisfy the voltage requirement. Then they will have to determine how the primary flux source will be generated (excitation), and how any required mechanical power can be transmitted into the electro-mechanical system, with the appropriate speed for the required frequency.
In all the above, we can have parallel paths of current, as well as of flux, in all sorts of combinations.

1) All ordinary AC power depends on electrical induction, which basically is flux variations through coils of wire. (In the stator windings).
2) Generator rotor current (also called excitation) is not directly related to Power Factor, but to the no-load voltage generated.
3) The reason for operating near unity Power Factor is rather that it gives the most power per ton of materials used in the generating system, and at the same time minimises the transmission losses.
4) Most Generating companies do charge larger users for MVAr, and for the private user, it is included in the tariff, based on some assumed average PF less than unity.
5) In some situations, synchronous generators has been used simply as VAr compensators, with zero power factor. They are much simpler to control than static VAr compensators, can be varied continuously, and do not generate harmonics. Unfortunately they have higher maintenance cost.
6) When the torque from the prime mover exceeds a certain limit, it can cause pole slip. The limit when that happens depends on the available flux (from excitation current), and stator current (from/to the connected load).

Induction machines testing

Case: We got by testing 3 different machines under no-load condition.
The 50 HP and 3 HP are the ones which behave abnormally when we apply 10% overvoltage. The third machine (7.5 HP) is a machine that reacts normally under the same condition.
What we mean by abnormal behavior is the input power of the machine that will increase dramatically under only 10% overvoltage which is not the case with most of the induction machines. This can be seen by the numbers given below.

50 HP, 575V
Under 10% overvoltage:
Friction & Windage Losses increase 0.2%
Core loss increases 102%
Stator Copper Loss increases 107%

3 HP, 208V
Under 10% overvoltage:
Friction & Windage Losses increase 8%
Core loss increases 34%
Stator Copper Loss increases 63%

7.5 HP, 460V
Under 10% overvoltage:
Friction & Windage Losses decrease 1%
Core loss increases 22%
Stator Copper Loss increases 31%

Till now, we couldn’t diagnose the exact reason that pushes those two machines to behave in such way.
Answer: A few other things I have not seen (yet) include the following:
1) Are the measurements of voltage and current being made by “true RMS” devices or not?
2) Actual measurements for both current and voltage should be taken simultaneously (with a “true RMS” device) for all phases.
3) Measurements of voltage and current should be taken at the motor terminals, not at the drive output.
4) Measurement of output waveform frequency (for each phase), and actual rotational speed of the motor shaft.

These should all be done at each point on the curve.

The reason for looking at the phase relationships of voltage and current is to ensure the incoming power is balanced. Even a small voltage imbalance (say, 3 percent) may result in a significant current imbalance (often 10 percent or more). This unbalanced supply will lead to increased (or at least unexpected) losses, even at relatively light loads. Also – the unbalance is more obvious at lightly loaded conditions.

As noted above, friction and windage losses are speed dependent: the “approximate” relationship is against square of speed.

Things to note about how the machine should perform under normal circumstances:
1. The flux densities in the magnetic circuit are going to increase proportionally with the voltage. This means +10% volts means +10% flux. However, the magnetizing current requirement varies more like the square of the voltage (+10% volt >> +18-20% mag amps).
2. Stator core loss is proportional to the square of the voltage (+10% V >> +20-25% kW).
3. Stator copper loss is proportional to the square of the current (+10% V >> +40-50% kW).
4. Rotor copper loss is independent of voltage change (+10% V >> +0 kW).
5. Assuming speed remains constant, friction and windage are unaffected (+10% V >> +0 kW). Note that with a change of 10% volts, it is highly likely that the speed WILL actually change!
6. Stator eddy loss is proportional to square of voltage (+10% V >> +20-25% kW). Note that stator eddy loss is often included as part of the “stray” calculation under IEEE 112. The other portions of the “stray” value are relatively independent of voltage.

Looking at your test results it would appear that the 50 HP machine is:
a) very highly saturated
b) has damaged/shorted laminations
c) has a different grade of electrical steel (compared to the other ratings)
d) has damaged stator windings (possibly from operation on the drive, particularly if it has a very high dv/dt and/or high common-mode voltage characteristic)
e) a combination of any/all of the above.

One last question – are all the machines rated for the same operating speed (measured in RPM

Active power losses in electrical motor

Equivalent active power losses during electrical motor’s testing in no-load conditions contain next losses:
1. active power losses in the copper of stator’s winding which are in direct relation with square of no-load current value: Pcus=3*Rs*I0s*I0s,

2. active power losses in ferromagnetic core which are in direct relation with frequency and degree of magnetic induction (which depends of voltage):
a) active power losses caused by eddy currents: Pec=kec*f*(B)x
b) active power losses caused by hysteresis: Ph=(kh*d*d*f*f*B*B)/ρ

3. mechanical power losses which are in direct relation with square of angular speed value: Pmech=Kmech*ωmech*ωmech,

Comment:
First, as you can see, active power losses in ferromagnetic core of electrical motor depend of voltage value and frequency, so by increasing voltage value you will get higher active power losses in ferromagnetic core of electrical motor.

Second, you can’t compare two electrical motors with different rated voltage and different rated power because active power losses in the ferromagnetic core, as I have already said above, depend of voltage value and frequency while active power losses in the copper of stator’s windings depend of square of no-load current value which is different for electrical motors with different rated power.

Third, when you want to compare active power losses in no-load conditions of two electrical motors with same rated voltage and rated power, you need to check design of both electrical motors because it is possible that one of them has different kind of winding, because, maybe in the past, one of them was damaged, so its windings had to be changed, what could be the reason for different electrical design and that has a consequence different no-load current value.